Resonant Power Converters with Switchable Resonant Modes

ABSTRACT

A system includes an input port having an input voltage, an output port having an output voltage, and a power converter having a switch network with a plurality of power switches and a first resonant tank having a first resonant capacitor and a first resonant inductor, where at least one resonant component within the first resonant capacitor and the first resonant inductor is a switchable component configured to switch between different values. The system further includes a resonant mode selection block configured to adjust a value of the switchable component to maintain a performance of the system, and a controller configured to adjust a switching frequency or a duty cycle of the power converter.

CROSS REFERENCE TO RELATED APPLICATIONS

This application is a continuation of U.S. patent application Ser. No.15/095,021, filed on Apr. 9, 2016, which is a continuation-in-part ofU.S. patent application Ser. No. 14/185,885 filed on Feb. 20, 2014, nowU.S. Pat. No. 9,344,042B2, which is related and claims priority to U.S.Provisional Application No. 61/850,972, titled, “High Efficiency PowerAmplifiers with Advanced Power Solutions” filed on Feb. 27, 2013, whichis herein incorporated by reference.

TECHNICAL FIELD

The present invention relates to power conversion and power amplifiers,and, in particular embodiments, to high efficiency power topologies andcontrol suitable for high efficiency power amplifiers and powerconverters.

BACKGROUND

Power amplifiers are widely used in wireless communication systems andother electronic devices, especially in mobile devices. To achieve highsystem efficiency and/or longer battery life, it is very important tomaintain high efficiency in power amplifiers.

The power amplifiers in many systems see signals with high peak toaverage power ratio. Because a typical power amplifier gets its energyfrom a power supply coupled to the drain (or the collector if atransistor is used) of its main power switch (a MOSFET or a transistor),its power efficiency can be improved by changing the drain voltage ofits main power switch through changing the voltage of its drain powersupply according to the envelope of the signal being processed by thepower amplifier. When the signal envelope has high bandwidth as oftenseen in today's wireless systems, changing the voltage of drain powersupply may cause the output signal of the power amplifier to bedistorted.

Moreover, as the bandwidth of the signal increases, the power supply'scontrol bandwidth should be increased accordingly. This put very highburden into the power supply, and the efficiency of the power supply islow with existing technology.

Improvements are needed to reduce the signal distortion in a poweramplifier with variable drain voltage, and to increase the efficiency ofthe drain power supply.

SUMMARY OF THE INVENTION

These and other problems are generally solved or circumvented, andtechnical advantages are generally achieved, by preferred embodiments ofthe present invention which provides an improved resonant powerconversion.

According to one embodiment of this disclosure, a system includes aninput port having an input voltage, an output port having an outputvoltage, and a power converter having a switch network with a pluralityof power switches and a first resonant tank having a first resonantcapacitor and a first resonant inductor, where at least one resonantcomponent within the first resonant capacitor and the first resonantinductor is a switchable component configured to switch betweendifferent values. The system further includes a resonant mode selectionblock configured to adjust a value of the switchable component tomaintain a performance of the system, and a controller configured toadjust a switching frequency or a duty cycle of the power converter.

According to another embodiment of this disclosure, an apparatusincludes an input port having an input voltage, an output port having anoutput voltage, and a transformer having a primary winding and asecondary wingding, where the primary wingding is coupled to the inputport. The apparatus further includes a power switch coupled between theprimary winding and the input port, a rectifier circuit coupled betweenthe secondary winding and the output port, a primary resonant tankhaving a primary resonant capacitor in parallel with the power switchand a primary resonant inductor including the primary winding, and asecondary resonant tank having a secondary resonant capacitor coupledbetween the secondary winding and the rectifier circuit, and a secondaryresonant inductor comprising the secondary winding, The apparatus alsoincludes a control block configured to control at least one of a dutycycle and a switching frequency of the power switch.

According to yet another embodiment of this disclosure, a methodincludes configuring a power converter comprising an input port havingan input voltage, an output port having an output voltage, a switchnetwork having a plurality of power switches, and a resonant tankcomprising a resonant capacitor and a resonant inductor, where at leastone resonant component within the resonant capacitor and the resonantinductor is a switchable component configured to switch betweendifferent values. The method further includes configuring a resonantmode selection block to adjust a value of the switchable component tomaintain a performance of the system, and configuring a controller toadjust a switching frequency or a duty cycle of the power converter.

The foregoing has outlined rather broadly the features and technicaladvantages of the present invention in order that the detaileddescription of the invention that follows may be better understood.Additional features and advantages of the invention will be describedhereinafter which form the subject of the claims of the invention. Itshould be appreciated by those skilled in the art that the conceptionand specific embodiment disclosed may be readily utilized as a basis formodifying or designing other structures or processes for carrying outthe same purposes of the present invention. It should also be realizedby those skilled in the art that such equivalent constructions do notdepart from the spirit and scope of the invention as set forth in theappended claims.

BRIEF DESCRIPTION OF THE DRAWINGS

For a more complete understanding of the present invention, and theadvantages thereof, reference is now made to the following descriptionstaken in conjunction with the accompanying drawings, in which:

FIG. 1 illustrates a schematic diagram of a power amplifier;

FIG. 2 illustrates a schematic diagram of a first illustrativeembodiment of a power amplifier in accordance with various embodimentsof the present disclosure;

FIG. 3 illustrates a detailed schematic diagram of an embodiment of apower converter in accordance with various embodiments of the presentdisclosure;

FIG. 4 illustrates a schematic diagram of an embodiment of a poweramplifier in accordance with various embodiments of the presentdisclosure;

FIG. 5 illustrates an embodiment of an envelope reference in accordancewith various embodiments of the present disclosure;

FIG. 6 illustrates an embodiment of a power amplifier in accordance withvarious embodiments of the present disclosure;

FIG. 7 illustrates an embodiment of a power amplifier with variousembodiments of the present disclosure;

FIG. 8 illustrates simulated waveforms of a power amplifier inaccordance with various embodiments of the present disclosure;

FIG. 9 illustrates an embodiment of a power supply architecture inaccordance with various embodiments of the present disclosure;

FIG. 10 illustrates embodiments of basic cells of an asymmetricmulti-level power synthesizer in accordance with various embodiments ofthe present disclosure;

FIG. 11(a) illustrates embodiments of an asymmetric multi-level powersynthesizer in accordance with various embodiments of the presentdisclosure;

FIG. 11(b) illustrates embodiments of an asymmetric multi-level powersynthesizer in accordance with various embodiments of the presentdisclosure;

FIG. 12 illustrates an embodiment of a multi-level power synthesizerwith linear control in accordance with various embodiments of thepresent disclosure;

FIG. 13 illustrates an embodiment of a multi-output power converter inaccordance with various embodiments of the present disclosure;

FIG. 14(a) illustrates an embodiment of a resonant power converter withautomatic frequency tracking in accordance with various embodiments ofthe present disclosure;

FIG. 14(b) illustrates simulated waveforms of the resonant powerconverter in FIG. 14(a) in accordance with various embodiments of thepresent disclosure;

FIG. 15(a) illustrates an embodiment of a resonant gate drive inaccordance with various embodiments of the present disclosure;

FIG. 15(b) illustrates the state-plane trajectory of the resonant gatedrive in FIG. 15(a) in accordance with various embodiments of thepresent disclosure;

FIG. 15(c) illustrates an embodiment of a resonant gate drive with gatevoltage shifting in accordance with various embodiments of the presentdisclosure;

FIG. 15(d) illustrates simulated waveforms of the resonant gate drive inFIG. 15(c) in accordance with various embodiments of the presentdisclosure;

FIG. 16 illustrates an embodiment of a resonant power converter withoutputs arranged in pairs in accordance with various embodiments of thepresent disclosure;

FIG. 17(a) illustrates an embodiment of a current-fed power converter inaccordance with various embodiments of the present disclosure;

FIG. 17(b) illustrates simulated waveforms of the power converter inFIG. 17(a) in accordance with various embodiments of the presentdisclosure;

FIG. 18 illustrates an embodiment of a multi-output power supply withvoltage regulation in accordance with various embodiments of the presentdisclosure;

FIG. 19 illustrates an embodiment of a multi-output current-fed fullbridge power converter in accordance with various embodiments of thepresent disclosure;

FIG. 20 illustrates simulated waveforms of the power converter in FIG.19 in accordance with various embodiments of the present disclosure;

FIG. 21(a) illustrates embodiments of a multi-output power converterbased on a buck topology in accordance with various embodiments of thepresent disclosure;

FIG. 21(b) illustrates embodiments of a multi-output power converterbased on a buck topology in accordance with various embodiments of thepresent disclosure;

FIG. 22 illustrates an embodiment of a dynamic power supply based on abuck topology in accordance with various embodiments of the presentdisclosure;

FIG. 23 illustrates an embodiment of a block diagram of a resonant powersupply in accordance with various embodiments of the present disclosure;

FIG. 24 illustrates an embodiment of a half-bridge resonant powerconverter in accordance with various embodiments of the presentdisclosure;

FIG. 25 illustrates an embodiment of a flyback resonant power converterin accordance with various embodiments of the present disclosure;

FIG. 26 illustrates an embodiment of a half-bridge resonant powerconverter in accordance with various embodiments of the presentdisclosure;

FIG. 27 illustrates an embodiment of a gate drive circuit for a powerconverter in accordance with various embodiments of the presentdisclosure; and

FIG. 28 illustrates an embodiment of a phase diagram of the gate drivecircuit shown in FIG. 27 in accordance with various embodiments of thepresent disclosure.

Corresponding numerals and symbols in the different figures generallyrefer to corresponding parts unless otherwise indicated. The figures aredrawn to clearly illustrate the relevant aspects of the variousembodiments and are not necessarily drawn to scale.

DETAILED DESCRIPTION OF ILLUSTRATIVE EMBODIMENTS

The making and using of the presently preferred embodiments arediscussed in detail below. It should be appreciated, however, that thepresent invention provides many applicable inventive concepts that canbe embodied in a wide variety of specific contexts. The specificembodiments discussed are merely illustrative of specific ways to makeand use the invention, and do not limit the scope of the invention.

The present invention will be described with respect to preferredembodiments in a specific context, namely in power amplifiers and powersupply technologies for power amplifiers. The invention may also beapplied, however, to a variety of other electronics systems, includingintegrated circuits, CPUs (central processing units), computers, telecomequipments, any combinations thereof and/or the like. Hereinafter,various embodiments will be explained in detail with reference to theaccompanying drawings.

Often a power amplifier is required to process a signal with a variableamplitude envelope which is related to the information to becommunicated. The envelope may have high-frequency components up toseveral tens or even hundreds of MHz in radio frequency (RF)communication systems and devices and other equipment. The carrierfrequency is usually much higher than the highest component frequency ofthe envelope. In the last few decades, people have tried to develop highefficiency technologies, such as envelope tracking (ET) and envelopeelimination and restoration (EER), for power amplifiers intended forvariable envelope applications. FIG. 1 shows a block diagram for such ahigh efficiency amplifier. S1 is the main switch. Although a MOSFET isshown as an example, it can also be a bipolar transistor (BJT) or otherswitching devices. L1 is the RF choke, C2 is the DC blocking capacitorto prevent dc component from getting into the RF output. C1 is thebypass capacitor to filter out noise in the power path. The enveloppower supply 101 provides drain power to the switch, and is designed tofollow the envelope information of the signal to be amplified. The biascircuit sets up the switch's operating point, and RF signal circuitconditions the signal to be amplified and couples it to the switch. Theimpedance match circuit 104 matches the impedance of S1 with theimpedance of the RF load connected to the power amplifier.

With EER technology (such as in polar modulation), S1 works in switchingmode (such as in Class E mode). The envelope power supply's output Vddis basically a replica of the envelope of the RF signal, and the RFsignal circuit for the gate (or base if a BJT is used) in the poweramplifier just processes the phase information of the RF input. If thetiming of the envelop signal is aligned perfectly with the phase signal,the RF output can have the right envelope and phase information. Becausein the switching mode the switch S1 works in two low loss states:saturation (ON) or block (OFF), the EER technique theoretically has highpower efficiency. However, it is extremely difficult to achieve a goodphase and envelope alignment, and it's very challenging to implement afast envelope power supply, so it's very expensive and difficult toimplement EER technology in practice.

With ET technology, S1 works in linear mode (such as in Class A, ClassAB or Class B mode). Vdd tracks but is slightly higher than theamplitude of RF output signal, and the gate signal is the same as in anormal linear amplifier (i.e. has both amplitude and phase information).With ET technology, the transistor's power loss is greatly reducedcompared to that in a normal linear power amplifier due to the reducedvoltage applied to the transistor. As a result, the ET power amplifieralso achieves high power efficiency. ET is a very promising technologyfor high efficiency linear power amplifiers, but similar to thelimitations of EER technology mentioned above, ET technology also facestremendous challenges:

The envelope power supply has to provide power with very wide frequencyspectrum. For example, in LTE technologies, the RF signal spectrum mayhave a bandwidth up to 20 MHz, but the envelope signal spectrum may havean effective bandwidth up to 100 MHz. The state of art power technologycannot process such wide-bandwidth power effectively;

The fast changing envelope power applied to the drain path of an ETpower amplifier causes significant RF signal distortion, and thustremendous predistortion needs to be used to regain the linearity of theamplification. Such predistortion normally requires a large amount ofcalibration work and complex procedure, and thus significantly increasesthe implementation cost.

The current ET technology doesn't consider the internal dynamics of apower amplifier to the drain power, and thus cannot be optimized forefficiency improvement reliably.

In this disclosure, new technologies addressing the above issues will beproposed and discussed in details. Although the description is made inthe context of ET technology, the principles can also be applied to EERtechnology and polar modulation, as they also involve interactionbetween the fast envelope power supply and the power amplifier itpowers, and the need of very fast dynamic power supplies.

FIG. 2 shows a simplified block diagram of a new ET+ technology.Although a single transistor Class A power amplifier is shown, similartechnologies can be applied to other power amplifier topologies, such asClass AB, Class B, Push-pull, Class E, Class F etc. Compared to thetopology in FIG. 1, the ET+ topology has an envelope interface circuit105 between the power switch and the optional impedance matching circuit104. In an embodiment in accordance with the present invention, theenvelope interface circuit 101 can be part of the impedance matchingcircuit 104. As well known in the industry, harmonic terminationfunction can be also part of the impedance matching circuit. Theenvelope interface circuit consists of an envelope frequency shunt 106and a carrier frequency path circuit 107. The circuit 106 presents a lowimpedance path to the envelope signal, thus allowing the envelopecurrent to flow through it and return to the envelope power supply 101with little impact to the output RF signal. The circuit 106 presents ahigh frequency to the carrier frequency, and thus doesn't impact RFsignal amplification significantly. On the contrary, the carrierfrequency path circuit 107 presents a high impedance path to theenvelope signal, forcing the envelope signal to go through circuit 106.To carrier frequency signal, the circuit 107 presents a low impedancepath, allowing the RF output signal to pass through it without muchdistortion or attenuation. With this architecture, clear envelope signalpath (with relatively lower frequency contents) and carrier frequencypath (with relatively higher frequency contents) are established. As aresult, the variable amplitude power applied to the drain path of thepower switch may not cause significant distortion to the RF output,significantly reducing the requirements of predistortion and itsassociated costs. This concept can be easily extended to other poweramplifier architectures, such as EER and polar modulation.

FIG. 3 shows a more detailed block diagram of the ET+ technology. Theenvelope interface circuit 106 is implemented with simple circuits. Inone embodiment, the envelope frequency shunt circuit 106 is implementedas an inductor L2 and an optional resistor R1. R1 also damps possibleresonance between reactive components, and so improves the signalquality in the amplifier. Moreover, R1 allows the drain voltage tochange relatively quickly with the change of the bias current or biasvoltage to the switch S1 when needed, and thus improves the envelopecontrollability of the architecture.

One of the key aspects to achieve high performance and high efficiencyin power amplifiers is to operate the switch at the right operatingpoint. As a minimum requirement, the valleys (the lowest points of thewaveform) of the drain voltage should be just slightly higher than zeroat any point of time, and the valleys of the source current should bejust slightly higher than zero under all conditions too. This requiresthe switch be biased according to the envelope of the RF input (oroutput) signal. As the envelope bandwidth increases over time for higherrates of information transfer, the drain voltage envelope of the switch(Vd) can have significant deviation from the power source voltage Vdddue to the complex dynamics of the power amplifier. As the envelopebandwidth is much lower than the carrier frequency, the envelopedynamics in FIGS. 3 and 4 is determined mainly by L1, C2, L2 and R1, andalso depends on the characteristics of the power switch S1. Propercontrol is needed to ensure the operating point of S1 to follow thesignal envelope properly. In FIG. 4, an envelope reference block 111converts the RF signal amplitude at input or output of the poweramplifier to envelope reference signals Vdr for the drain voltage andIdr for the source current of S1. The reference is based on the signalenvelope, and a buffer signal is added to the signal envelope to makesure that the envelope reference is higher than the required signalenvelope by a certain margin, as is exemplified by FIG. 5. The buffervalue can be fixed or adjusted according to actual operation so that theperformance can be optimized. Also, the reference signals may not havethe exact replica of the envelope signal, and some of the highestfrequency components in the envelope signal may be reduced or removedfrom the reference signals to reduce implementation complexity and cost.An envelope detect circuit 113 is coupled to the drain of S1, so itsvoltage envelope Vd is sensed and used as a feedback signal. Similarly,the source current of S1 is sensed and coupled to an envelope detectcircuit 114 to generate the envelope signal Is of the source current. Acontrol block 112 may consist of feed-forward function, feedbackfunction and any combination thereof. The block 112 processes thereference signals Vdr, Isr and feedback signals Vd and Is, to generatethe drain power reference Vddr, and bias control reference Vc. Thedynamic power supply 101 generates a drain power Vdd according to Vddr,while the bias control circuit 102 uses Isr as the reference to controlthe bias of S1. The RF signal conditioning and control, which mayinvolve functions such as amplification, delay, filtering, alignment etcare performed by the RF signal circuit and are not discussed in detailshere, as they do not directly contribute to the envelope control.Although voltage signals are used to describe the envelope signal Vd, Isand their references, some or all of these signals can also be currentsignals.

The control architecture in FIG. 4 can have many differentimplementations. FIG. 6 shows a preferred embodiment, in which both thedrain voltage envelope and source current envelope are controlledthrough feedback control. A difference block generates the drain voltageenvelope error signal by comparing the reference signal Vdr and theenvelope feedback signal Vd. Then the error signal is processed bycompensation block 115 to generate the drain power reference Vddr.Another difference block generates the source current envelope errorsignal by comparing the reference signal Isr and the envelope feedbacksignal Is. Then this error signal is processed by compensation block 117to generate the bias control voltage Vc. FIG. 7 shows another preferredembodiment in which both feedback control and feed-forward control areused to accomplish the envelope control function. In accordance withthis embodiment, a feed-forward control block 118 is used to process thereference signal Vdr. Such feed-forward function can consider knowneffects of the power amplifier dynamics. However, due to parametervariations and other deviation from ideal condition, it may not bepractical and economical to rely on the feed-forward control alone forthe drain voltage envelope control. Therefore, a feedback control block116 similar to the one in FIG. 6 is also included to fine tune thecontrol action for the drain voltage. The outputs of the block 118 andthe block 116 are summed together to adjust the output of the dynamicpower supply 101. Because of the added feed-forward control, theoperating range and requirements of the feedback control in FIG. 7 canbe lower than that in FIG. 6. Since the low-frequency dynamics betweenthe source current and gate voltage of a switch is usually simpler thanthe relationship between the drain voltage and power input, it'spossible to use just feed-forward control for the source currentenvelope control, as is shown in FIG. 7 (through feedforward block 119).However, if the effect of the drain voltage on the gate voltage (theMiller effect) is to be considered, an optional link can be establishedby sending the drain envelope signal Vd into the feed-forward block forthe source current control, so proper measures can be taken to reduce oreliminate the miller effect in the source current.

FIG. 8 shows simulated waveforms of a drain voltage envelope and itsreference in an embodiment. Clearly, with proper design the drainvoltage envelope can be controlled to follow its reference closely.

A key aspect of the above implementation is to implement ahigh-efficiency dynamic power supply with very wide bandwidth.Generally, to achieve a high efficiency power conversion, a switchingpower converter is preferred over a linear power supply. However, astate of art switching power converter may not be able to have wideenough bandwidth. Some people proposed to use a combination of switchingpower converter and a linear power supply in a series or parallelconfiguration. Such combination naturally has relatively low efficiencyand high control complexity. Another proposed method is to use amulti-output power supply followed by a voltage selection circuit, andthe voltage selection circuit selects one of the voltages as the outputof the dynamic power supply. However, to achieve a good result, thenumber of the voltages should be high, thus resulting in highimplementation cost.

A new power architecture is shown in FIG. 9 for fast dynamic powersupplies. A multi-output power supply 120 with a limited number ofdifferent output voltage levels are followed by an asymmetricmulti-level power synthesizer 130. The power synthesizer 130 adds itsinput voltages in different ways so that its output voltage has a muchhigher number of voltage levels than its number of input voltages.

FIG. 10 shows the basic cells of an asymmetric multi-level synthesizer.FIG. 10(a) is a two-level cell. Its output is zero when S1 is ON, and V1when S2 is ON. S1 and S2 are controlled to be in complementary mode witha short transition between states in which both switches are off toavoid cross conduction. FIG. 10(b) shows a four-level cell, with anasymmetric input voltage pair 152, and a four-switch network 151consisting of a switch pair S1 and S2 in a totem pole configuration, anda second switch pair S3 and S4 in another totem pole configuration. Anoutput terminal is coupled to the common node of each totem pole (i.e.at the junction of the top switch and the bottom switch in the pair).Two voltage sources at the input can have a substantially doublingrelationship as described above: V2=2*V1. There are 4 possible states:

S1 and S4 OFF, S2 and S3 ON: Vo=0;

S1 and S3 ON, S2 and S4 OFF: Vo=V1;

S2 and S4 ON, S1 and S3 OFF: Vo=V2=2V1;

S1 and S4 ON, S2 and S3 OFF: Vo=V1+V2=3V1

Again, the S1 and S2 switch pair, as well as the S3 and S4 switch pair,are controlled to be in complementary mode so at most one switchconducts in a pair at any time. There is a short transition period ifthe switches in a pair need to change states during which both switchesare off to avoid cross conduction. With this structure, if the inputvoltages of synthesizer have a doubling relationship between them (suchas V2=2*V1, V3=2*V2 and so on), the synthesizer basically becomes abinary encoder of voltages (power), and the levels of the output voltagehave even step size between them. For example, if the multiple-outputpower supply's output includes 1V and 2V, then the output of thesynthesizer can be 0V, 1V, 2V, and 3V. Generally, n voltages at theinput of the synthesizer can be synthesized into 2^(N) different voltagelevels at the output using 2N switches. However, in other embodimentsthe input voltages may not follow the doubling relationship preciselyand the step size between adjacent voltage levels does not have to bethe same in the whole range. Generally, it is desired that the doublingrelationship substantially to achieve good performance. For example, thehigher voltage in a four-level cell should be maintained within a rangeof 1.5 to 2.5 times the lower voltage. Although the doublingrelationship will be used in the following description, the principlesof power conversion, power synthesizer and control can be easilymodified for other voltage relationships.

Any number of two-level cells and four-level cells can be connected inseries by putting their output voltages in series to get a higher numberof voltage levels. For example, a configuration with a two-level cell inseries with a four-level cell makes an 8-level synthesizer, and aconfiguration with a four-level cell directly in series with anotherfour-level cell makes a 16-level synthesizer. Note that the inputvoltages of the adjacent cells should also follow a voltage doublingrelationship, i.e. the lower input voltage in the cell with highervoltages should be twice the higher input voltage in the cell with lowervoltages. More voltage levels can be obtained by putting more cells inseries, as this structure is easily expandable. Generally, a switchnetwork with a N1-switch cell and a N2-switch cell is capable ofproducing N1 times N2 voltage levels. When one more cell with N3switches is added, the number of voltage levels the switch network canproduce is increased to N3 times the original number of voltage levels.Also, if a switch network has N input voltages, it is capable ofproducing 2^(N) voltage levels using 2N switches. FIG. 11(a) showsexamples of an 8-level synthesizer, and FIG. 11(b) shows a 16-levelsynthesizer. The impedance network 141 is coupled between the switchnetworks and the output and reduces the current flowing through theswitches during the switching transition between different states, andthus increases the efficiency. The impedance network has naturallyfiltering function working together with the output capacitor Co. In onepreferred embodiment, the impedance network is an inductor, names as L1.L1 and the output capacitor Co may become part of a resonant tank. Thevalue of L1 should not be too high to avoid significant voltageoscillation at the output. In a preferred embodiment, the characteristicimpedance

${{Zo} = {\sqrt{\frac{L\; 1}{Co}} \approx {Rs}}},$

where Rs is the resistance in the current path inside the powersynthesizer, including the switch resistances, connection resistance,and resistance of L1. This means the resonant tank consisting of L1 andC1, as well as Rs has a quality factor (defined as Zo/Rs) ofsubstantially close to 1, for example in the range of 0.4 to 2.5. L1 canbe a discrete inductor, or the parasitic inductance in the current pathof the power synthesizer including the package inductances of theswitches, and any combination thereof. In a preferred embodiment, Co canbe placed at or inside the power amplifier the synthesizer powers. Ifthe synthesizer and the power amplifier are in different dies ordifferent packages, the parasitic inductance from the packages andinterconnection between them may be enough to serve the purpose of L1.

As explained above, an asymmetric power synthesizer behaves like a powerencoder. The drive signals for the switches in a power synthesizer canbe obtained from a decoding process of the digital representative of thedesired voltage. If the desired voltage is not in a digital formatoriginally, an ADC converter can be used to digitize the required value.This control scheme is especially suitable for digital implementation.

The output voltage of a power synthesizer as described above hasdiscrete values as it changes from one level to another level rapidly.Sometimes it's desirable to have a smoother and more accurate output. Alinear regulator can be added at the output to achieve the smoothingfunction during a level change and/or to realize accurate voltagecontrol. This linear regulation can also be implemented by controllingthe gate drive voltage amplitude of switches within the synthesizer.FIG. 12 shows an example, with a four-level symmetric cell 150 andanother four-level cell 151, and an impedance cell 141, and outputvoltage C0. Again, the impedance network 141 reduces the current flowingthrough the switches during the switching transition between differentstates, and thus increases the efficiency. The impedance network hasnaturally filtering function working together with the output capacitorCo. In one preferred embodiment, the impedance network is an inductor,names as L1. L1 and the output capacitor Co may become part of aresonant tank. A control circuit 142 is used to adjust the amplitude ofthe driving voltage of S1 and S2. The timing of S1 and S2's gate driveis still controlled by the decoder discussed above, and other switchesare still controlled by the decoder. The linear control of S1's and S2'sdrive voltage changes the resistance of this switch pair, and thus canachieve fine-tune regulation of the output voltage of the powersynthesizer. This linear control can also be used to reduce the outputoscillation if needed. Obviously, the range of the linear adjustment ofthe output voltage can be less than 1 step size of the voltage levels,and thus high efficiency can still be achieved. This is a simple and lowcost implementation of accurate and fast dynamic control in dynamicpower supplies.

The power synthesizer needs multi voltage rails at the input. Thesevoltage rails can be generated by multi-output power supplies. Suchpower supplies can be implemented with different topologies, such asflyback converters, forward converters, etc. FIG. 13 shows anarrangement with a flyback topology with 2 pairs of output voltages (171and 172, more outputs can be added easily by repeating the pattern). Ifthe relationship between the number of turns of the secondary windingshas a doubling relationship:

n2=2*n1,n3=2*n2,n4=2*n3

Then ideally the output voltage will also have a doubling relationship:

V2=2*V1,V3=2*V2,V4=2*V3

By controlling of duty cycle of the primary switch Sp1, the outputvoltages can be regulated to the desired value. S1 through S4 in thesecondary are synchronous rectifiers and can be replaced with diodes ifdesired. The output voltages can be easily arranged into asymmetricvoltage pairs (171 and 172) to suit the needs of a power synthesizer.

In some applications, the input voltage Vin may already be at the rightvalue for the system, so the multi-output power supply just needs toconvert the input voltage to needed multi-output format, and does notneed to adjust values of its outputs. In this case the multi-outputpower supply behaves as a multi-output bus converter. FIG. 14(a) shows amulti-output resonant converter with local compensation suitable forsuch applications. Half-bridge configurations at the primary side (180)and at the secondary side (181 and 182) are shown in the figure.However, as well known in the industry, full-bridge and other suitableconfigurations, and any combination thereof, can also be used.Traditionally, a resonant converter such as an LLC converter has aresonant tank only at the primary side of the transformer. When such atopology is extended to multi-output configuration by adding moretransformer windings, each output's voltage is affected by the leakageinductance of the secondary winding to which it is coupled. Therefore,the voltage of the each output may deviate significantly from the idealvalue determined by the turns-ratio of the transformer windings,resulting in voltage inaccuracy. In the new topology of FIG. 14, eachoutput also has a resonant tank coupled to the associated secondarywinding, consisting of two resonant capacitors in half-bridgeconfiguration (e.g. Cr11 and Cr12 for V1) and a resonant inductor (e.g.Lr1 for V1). This double-side resonance topology significantly reducesthe effect of transformer leakage inductance on voltage regulation, andthus achieves better cross regulation between different outputs, andalso allows the output voltage to be sensed at the primary side moreaccurately (from feedback winding 161, for example). The resonantinductors can be discrete parts, or the leakage inductance of thetransformer windings, or any combination thereof. Series resonant tanksare shown in the figure, but other types of resonant tanks can also beused, including LLC (utilizing the magnetizing inductance of thetransformer), LCC, parallel resonant, series-parallel resonant,multi-element resonant etc. However, the resonant tanks at the primaryside and the secondary side should have the same configuration, and thecomponents in the resonant tanks should have about the same equivalentvalues (considering the scaling factor of transformer's turns ratios).All the resonant tanks should be designed to have the same orapproximately the same resonant frequency (or resonant frequencies ifmore than one resonant frequencies existing in a tank). If theconverter's switching frequency is the same as the designed resonantfrequency, and thus the leakage inductance (or any added inductance tothe resonant tanks) is compensated by the associated resonantcapacitors, and resonant tanks have a voltage gain of 1. Under thiscondition, the output voltages follow the exact relationship determinedby the transformer winding turns-ratios, if the power losses of thecomponents are ignored:

${{V\; 2} = {\frac{n\; 2}{n\; 1}V\; 1}},$

where n2 and n1 are the number of turns of the secondary windingscoupled to V2 and V1 respectively.

Even if the operation frequency is not exactly the same as the resonantfrequency of each resonant tank due to component tolerance and otherissues, the compensation effect of the resonant capacitors can stillimprove the performance of the power supply. The feedback winding 161(with of turns) picks up a reflected voltage whose amplitude is inproportion to the secondary winding voltages of the transformer. Afterprocessed by the signal conditioning block 162, the resulting feedbacksignal coupled to the error amplifier (E/A) is proportional to theoutput voltages. In one preferred embodiment, the functions performed bythe block 162 include rectification and filtering, and possibly somesignal conditioning to remove the effect of noise and irregularitycaused by switching actions, By making the error/amplifier's referencevoltage proportional to the input voltage with the right scaling factorM, all resonant tanks should have a theoretic voltage gain of close toone, so the effective voltage gain of the equivalent resonant tankrepresenting the multi-tank system in this converter is approximately 1in steady state. Considering the parameter tolerance in the resonanttanks, the reference, and the feedback control, the steady-stateswitching frequency may not be exactly the intended resonant frequencyof the resonant tanks, or the same as the actual resonant frequency ofany resonant tank. However, it will not deviate significantly from theintended resonant frequency if the component tolerance is not too high,for example not higher than +/−20%. This mode of operation is calledcontrolled near resonance operation. As a result, the converter willautomatically operate at a frequency very close to (for example within10%) the resonant frequencies of all resonant tanks, achieving goodefficiency and good output voltage regulation without sensing eachoutput. Please note that in other embodiments this control scheme canwork well also by sensing the output voltage directly at one of theoutputs (instead of through a feedback winding), but such control systemwith direct sensing is more complex and expensive due to the need ofsignal isolation.

The existence of a closed-control loop in the controlled near resonanceoperation also allows the transient performance be controlled ifnecessary. Especially, for large transients such as startup or faultprotection, a soft start or soft transition can be controlled throughduty cycle or switching frequency of the converter, or throughtrajectory control of the resonant tank variables (the current in aresonant inductor, transformer winding or switch, the voltage across aresonant capacitor, or both).

FIG. 14(b) shows the simulated waveforms of the converter in FIG. 14(a)in steady state operation. The primary switches Sp1 and Sp2 are gated onand off in complementary mode, with each having about 50% duty cycle. Ashort transition period is added between a switch's turn-off and theother switch's turn-on, to avoid cross conduction and allow zero-voltageturn on. It can be seen that zero voltage turn-on has been achieved forboth Sp1 and Sp2. However, in other embodiments the resonant convertercan be operated to achieve zero-current turn-off of Sp1 and Sp2, as iswell known in the industry. Usually, to achieve zero voltage turn-on themagnetizing inductance of the transformer T is intentionally reduced. Itcan been seen also that the amplitude of the reflected voltage from thefeedback winding nf is very close to an output voltage with same numberof turns as nf. The small difference is mainly due to the power lossesin the output circuit. Such deviation can be reduced if desired byconsidering power loss related information such as the transformer orswitch current in the feedback or reference circuit of the erroramplifier denoted as E/A in the regulation circuit 165. Althoughsynchronous rectifiers S1 through S4 are shown in the secondary in FIG.14(a), diode rectifiers can also be used to replace any or all of thesynchronous rectifiers, as usual in power converters. The gate drivesignals of synchronous rectifiers can be obtained from control signal inthe primary through a gate drive transformer or other isolationcomponents, or from windings on the main power transformer T, or fromthe current signals in the associated secondary windings or from thecurrents in the devices themselves. More outputs can be obtained byadding more secondary windings and associated resonant tanks andrectifier circuits.

This topology can also be used to operate away from the resonantfrequency and thus achieves different voltage gains to regulating theoutput voltages to desired values. As long as the equivalent LC valuesin each resonant tank (considering the scaling factor of thetransformer's turns ratios) are approximately the same, the double sideresonance can achieve good cross regulation between the outputs.

The main power switches (usually power MOSFETs) should be designed tohave low gate charge, so the driving loss is not significant. This meansthat the gate threshold voltage of the switch and the drive voltageapplied on the gate should be reduced whenever possible. If theswitching frequency used is high, it may be desirable to use resonantgate drive technologies to further improve the power converterefficiency. For a converter running around resonant frequency, it's alsopossible to use similar timing and drive circuit of the primary powerswitches (in this example Sp1 and Sp2) to drive the synchronousrectifiers, making the converter design much easier. FIG. 15(a) shows anexample of a resonant gate drive circuit. A bridge circuit consisting ofdrive switch Network 180 (S1 through S4) converts the control signal fora controller to an ac waveform. The ac waveform is fed into a resonantcircuit coupled to a gate drive transformer T1. Cp in the primarywinding of T1 can serve for dc-blocking, or be part of the resonantcapacitance, or be used for both purposes. Lp, Ls1, Ls2, and Lsx are theresonant inductors, and each of them can be a discrete inductor, theleakage inductance of the corresponding transformer winding, or anycombination thereof. Cg1, Cg2 and Cgx are the equivalent gate-sourcecapacitance of the switches to be driven, or any additional capacitanceput there, or any combination thereof. Cg1, Cg2, and Cgx are also partof the resonant capacitance. To make sure that each drive output hassimilar performance, it's important to make sure that each componentassociated with each drive winding of transformer (n1, n2, nx) hassimilar equivalent value to its counterpart in any other driver windingwhen transferred to the primary side. If different numbers of turns areused in the driving windings, then in any secondary winding circuit, thesecondary inductance (Ls1, Ls2, and Lsx) should be approximatelyproportional to the square of the number of turns of the winding, whilethe capacitance should be approximately inversely proportional to thesquare of the number of turns of the winding. If the equivalentgate-source capacitance of any switch is different from another's,external capacitors of appropriate values can be used to make theequivalent capacitance reflected to the primary approximately the samefor all drive windings. In this way, all the resonant components can begrouped into a series resonant tank of inductance Lr (including theleakage inductance of the transformer and any trace and packaginginductance) and resonant capacitance Cr. Traditionally, a resonant gatedrive involves one resonant state in one direction (charging anddischarging), so the gate voltage is basically sinusoidal. Issues withthis kind of driving scheme is that it takes a relatively long time fora switch to change from OFF state to ON state or vice versa, and thegate drive waveform varies strongly with the variation of the inductanceand capacitance values. As a result, the converter performance iscompromised. To improve upon this, a new composite driving scheme can beused. FIG. 15(b) shows the concept of the composite driving technique instate plane. As is well known, a resonant tank's normalized inductorcurrent i(Lr) and the normalized capacitance voltage moves in a circularfashion in a resonant state. With the composite driving technology, theresonant tank operates in more than one state in the charging phase anddischarging phase of a switching cycle.

In FIG. 15(b) two states are shown for each direction. The first stateis shown by the big arcs in the trajectory, during which the voltageapplied to the resonant tank is zero, and the resonant voltage (and thusgate voltages) changes polarity, as the center of the arcs is (0, 0).This can be called the polarity reversing state. The other state isshown by the small circles, during which the drive power supply Vcc isapplied to the resonant tank in positive or negative direction, so thecenters of the circles are (+Vcc, 0) and (−Vcc, 0) respectively. Becausethe radius of the small circles is much smaller than Vcc, during secondstate the resonant voltage fluctuate around Vcc or −Vcc slightly, soroughly the gate voltage can be considered to be pseudo “clamped” aroundVcc or −Vcc with a small ripple. This state can be called a pseudoclamped state. Because a trajectory can circle the center in secondstate (the pseudo clamp state) as many times as required, the gate drivevoltage can stay in the pseudo “clamped” mode for a relatively long andcontrollable time which is terminated by the start of the polarityreversing state in the other direction.

A complete switching cycle can be divided into 4 phases. The followingis an example for explanation:

Assuming at the start the drive circuit is in positive charging phasewith S2 and S4 turned on (S1 and S3 are off), so the circuit is inpositive pseudo clamp state. The drive outputs Drive 1 and Drive X havepositive voltages around a maximum value determined by Vcc with ascaling factor associated with the winding turns ratios and the ratiosof resonant capacitances. The switches coupled to Drive 1 and Drive Xare turned on. The other drive output Drive 2 has a voltage aroundnegative maximum, so the switch coupled to it is off. The drive circuitand the switches will be in this state until S2 is turned off and S1 isturned on.

After S2 is turned off and S1 is turned on, the drive circuit entersinto negative polarity reversing state, and the resonant voltage andthus the gate voltages cross zero and reverse their polarities.

When the resonant capacitor voltage reaches close to −Vcc (or thecurrent in Lr is about zero), S4 is turned off and S3 is turned on, nowthe drive circuit enters the negative pseudo clamp state, and the gatevoltages are in pseudo clamped mode again, but the voltage polaritiesare versed compared to the starting point mentioned above. This stateterminates until S1 is turned off and S2 is turned on.

After S1 is turned off and S2 is turned on, the resonant voltage andgate drive voltages start their journey to change polarity again inpositive polarity reversing state.

When the resonant voltage V(Cr) reaches close to Vcc, S3 is turned offand S4 is turned on, the drive circuit enters into the positive pseudo“clamp” state, and the system finishes a full switching cycle.

By adjusting the relative duration of the pseudo clamp mode, the drivecircuit's operation is more independent of the resonant tank parameters,and the transition time between a switch's states can be faster than intraditional control. Moreover, the duty cycle of the main switches caneven be adjusted as the durations of pseudo clamp modes do not have tobe equal in positive direction and negative direction. This gives betterperformance and freedom in the converter design. The key to this schemeis that the control timing of the drive switches S1 through S4 should bedetermined by comparing the resonant capacitor voltage to Vcc and −Vcc,or comparing the resonant inductor current to zero (or a small value),so this is basically a drive scheme with feedback. Please note thatactual feedback signal, either the resonant capacitor voltage or theinductor current, doesn't have to be measured at the primary side of thetransformer T1. As long as the proper scaling is considered (due to thetransformer turns ratios and the LC component values), the feedbacksignal can be taken from one or more driving circuit on the secondaryside of the transformer, or from one or two feedback windings on thetransformer. Alternatively, the duration of the polarity reversingstates can be fixed according to the resonant tank LC parameters(roughly half the resonant cycle). This is a simple open loop control,but should work reasonably well for many applications. Also, if moredrives are needed, more driving windings and associated circuit can beadded as this drive structure is easily expandable.

The above schemes are relatively easy to implement with modern analogand/or digital electronics. The above explanation is for steady stateoperation. During transients, such as during a startup, the controlshould be slightly modified. For example, during a startup, the switchfrequency and/or duty cycle should be changed to reduce voltage andcurrent stresses in both the drive circuit and the main power circuit.One possibility is to keep the duration of the polarity reversing stateabout the same as in steady state (without comparing the capacitorvoltage), but gradually increase the duration of pseudo clamp statesfrom zero to normal value in a soft-start process. Other control methodssuch as switching frequency control, duty cycle control, and trajectorycontrol can also be used to limit the stress level. For faultprotection, hick-up schemes with shutdown and soft-start phases can beconsidered.

The above circuit many have a draw back for some applications: a gatedrive voltage swings to both positive and negative value symmetrically,and its dc component is zero. It may be desirable to move the gate drivevoltage to more positive or more negative in certain applications, forpurposes such as adjusting the dead-time between switches. FIG. 15 (c)shows an implementation with voltage shifting and voltage clampingtechniques. Diodes D1 and D2 create a positive dc voltage for Drive 1and Drive 2, so the voltage waveforms will be shifted up for both driveoutputs. If a negative bias is desired in any output, the polarity ofthe associate diode can be reversed. The optional resistors R1 and R2are used to discharge the gate voltages. By adjusting the values of Cs1and Cg1, the waveform of Drive 1 voltage can be shifted up and down.Similarly, the voltage of Drive 2 can be shifted by the values of Cs2and Cg2. The switch Sg3 clamps the negative voltage of Drive 3 to aboutzero. When the clamp switch is turned on, the resonant capacitor itclamps is shorted, and totally resonant capacitance of the resonant tankis changed. As a result, the voltage waveform may be changed. Thecapacitances of Cs3 and Cg3 and the number of turns of winding n3 can beoptimized so the effect of this capacitance change is reduced. As analternative, if needed another clamp switch can be put in parallel withCs3, with its gate connected at the drain of Sg3. With the symmetricconnection of these two switches, if Cs3 and Cg3 have the samecapacitance, then the equivalent capacitance in the resonant tankdoesn't see much change during the different phases of operation, andthus the gate voltage waveform is more consistent. Such a symmetricarrangement of clamp switches is shown with the circuit coupled towinding nx. The driving winding nx in this figure powers two driveoutputs coupled together. These two outputs have reverse polarities, andcan be used to drive for example two bottom switches in a bridgeconfiguration or the power switches in a push-pull configuration. Theoptional clamp switches Sgx1 and Sgx2 can clamp the negative gatevoltages to zero if needed. As the clamp switches is turned onalternatively, the resonance of this resonant tank is not affected muchby the voltage clamp. More drive outputs can be added by repeating thecircuit patterns in FIG. 15(c), and the voltage shifting and voltageclamping techniques shown in the figure can be used in any combinationas needed.

FIG. 15(d) shows key simulated waveforms of the circuit shown in FIG.15(c), and verifies the operation principles discussed above.

The above discussed gate drive technologies can be used to drive powerswitches and synchronous rectifiers in different topologies. Thecontroller can take the feedback information of resonant capacitorvoltage or transformer current to accomplish closed-loop driving, or usea configurable delay corresponding to the polarity reverse state toaccomplish open loop driving.

The topology in FIG. 14(a) uses one driving winding for each outputvoltage. To simplify the implementation, the output voltages may becombined into voltage pairs as needed by a power synthesizer. FIG. 16shows a practical implementation. Now the resonant capacitors in thesecondary side of the transformer Cr1, Cr2, cr3 and Cr4 are arranged tobe in series with the secondary windings, and will resonate with theleakage inductance of their respective secondary windings. Since theleakage inductance of a secondary winding is proportional to the squareof the number of turns of the winding, the resonant capacitance coupledto each winding should have a reciprocal relationship to the square ofthe number of turns of the winding so that all the resonant tanks canresonate at the same frequency, i.e.:

${{{Cr}\; 2} \approx {\frac{n\; 1^{2}}{n\; 2^{2}}{Cr}\; 1}};\mspace{14mu} {{{Cr}\; 3} \approx {\frac{n\; 2^{2}}{n\; 3^{2}}{Cr}\; 2}};\mspace{14mu} {{{Cr}\; 4} \approx {\frac{n\; 3^{2}}{n\; 4^{2}}{Cr}\; 3}};$

If needed, discrete resonant inductors can be put in series with thetransformer windings, and the value of resonant capacitors should beadjusted accordingly.

To simplify the transformer structure, the secondary windings in eachvoltage pair circuit is connected together: n2 is connected with n1,while n4 is connected with n3. Please note that the polarity of n2 isreverse to n1's polarity, and the polarity of n4 is reverse to n3's.This reverse arrangement of windings allows the rectifiers (synchronousrectifiers or diodes) to be connected together in a dual totem network181 and 182, as is shown in FIG. 16. The outputs are arranged inasymmetric pairs 191 and 192. With this arrangement, when Sp1 conducts,synchronous rectifiers S11, S22, S31 and S42 should be gated on. WhenSp2 conducts, synchronous rectifiers S12, S21, S32, and S41 should begated on. Because now the top winding in a voltage pair circuit chargesboth output voltages in the pair, to maintain a doubling relationshipbetween the output voltages in the pair, the top winding should havethree times the number of turns as the bottom winding: n2=3*n1, n3=4*n1,n4=3*n3.

The doubling relationship of output voltages can also be realized withsimpler winding structure by duty cycle control. In the half-bridgetopology shown in FIG. 17(a), multiple windings are coupled to the samemagnetic component (transformer T1 in this case), where there exists arelationship between the numbers of turns in the windings: n3=4*n2. Sp1and Sp2 are controlled in complementary mode but with different dutycycles. The duty cycle of Sp1, D1, is twice the duty cycle of Sp2, D2:D1=2*D2. If the short transition periods in which both switches are offare ignored, then we have:

${D\; 1} \approx {\frac{2}{3}\mspace{14mu} {and}\mspace{14mu} D\; 2} \approx {\frac{1}{3}.}$

Then by virtue of volts-second balance of the magnetic component T1, theoutput voltages in steady state have a doubling relationship:

${{V\; 1} = {{Vin}*\frac{n\; 2}{3*N\; 1}}};{{V\; 2} = {2^{*}V\; 1}};{{V\; 3} = {4^{*}V\; 1}};{{V\; 4} = {8^{*}V\; 1}}$

In this way, one secondary winding is used to produce a pair of outputs.In one embodiment, the two output voltages are different in a pair. Inother embodiments the two voltages can be made to have a differentrelationship by adjusting D1 and D2 differently. The synchronousrectifiers S1 through S4 can be replaced by diodes if needed. The gatedrive signals for the synchronous rectifiers can be in synch with theprimary switches, or emulate ideal diodes: gated on when the body diodestarts conducting, and gated off when the current through it is lowerthan a threshold value. The drive signals can be obtained from primarycontrol signal or through transformer windings of T1, or through currentsignals in the transformer windings or the switches. As an option, theabove discussed composite resonant driving technology can also be used,as the durations of the pseudo clamp state can be set to differentvalues during Sp1 conducting phase and Sp2 conducting phases.

In practice, closed-loop control may be required to maintain the voltagerelationship. A possibility is to use voltages in the primary side, suchas capacitor voltages or a winding voltage to simplify the control cost.For example, when the voltage of Cp1 is less than 50% of the voltageacross Cp2 (or less than ⅓ of the dc-link voltage across the half-bridgeconfiguration of Sp1 and Sp2), then the duty cycle of Sp1 (D1) should beadjusted lower. When the voltage of Cp1 is higher than 50% of thevoltage across Cp2, then the duty cycle of Sp1 (D1) should be adjustedhigher. A feedback controller can automatically adjust the duty cycle tothe right value.

Due to current-fed nature of this topology caused by the input inductorL1, a snubber circuit or clamp circuit may be used to reduce the voltagestress of primary switches. In FIG. 17(a) an optional passive snubberconsisting of Cd and Rd is inserted across the dc link and between theinput inductor L1 and the switches. Other types of snubber or clampcircuits can also be used. FIG. 17(b) shows simulated waveforms of thistopology. It can be seen that Sp1 is turned on with reduced voltage, andSp2 is turned on with zero voltage. The ability to achievesoft-switching in this topology improves its efficiency.

The topology in FIG. 17(a) has limited ability to adjust the outputvalues by adjusting the duty cycles of the switches. If a wide range ofadjustment is needed, a regulation stage can be added at the input. FIG.18 shows a buck regulation stage 195 consisting of buck switch Sb1 andsynchronous rectifier Sd1 is added. The switching of Sb1 can besynchronized with Sp1 and Sp2, or can be arranged independently.

Different implementations can be obtained by changing the half-bridgeconfiguration in FIG. 17(a) to different topologies. A full-bridgeconfiguration using the same duty cycle control principle is shown inFIG. 19 which can two pairs of outputs. The secondary windings also havea quadruple relationship between the number of turns: n3=4*n2. Theprimary switch network consisting of four primary switches Sp1 throughSp4 are operated in 3 states alternatively:

State 1: Sp1 and Sp3 are turned on, and Sp2 and Sp4 are turned off.Secondary side synchronous rectifiers S1 and S3 conducts, and S2 and S4are gated off. The transformer winding voltages are determined by V1 andV3;

State 2: Sp1 and Sp3 are turned off, and Sp2 and Sp4 are turned on.Secondary side synchronous rectifiers S1 and S3 are gated off, and S2and S4 are gated on. The transformer winding voltages are determined byV2 and V4;

State 3: all primary switches Sp1 through Sp4 are turned on (or at leastswitches in one of the two legs, i.e. Sp1 and Sp2 or Sp3 and Sp4, areturned on), all secondary synchronous rectifiers S1 through S4 are gatedoff. The transformer winding voltages are zero.

In a switching cycle these states may have different sequences, but apreferred sequence is that an active state (State 1 or State 2) isfollowed by a zero state (State 3), and then the other active state(State 2 or State 1), and then a zero state (State 3). The duty cycle ofState 1 (D1), should be kept at twice the duty cycle of State 2 (D2):D1=2*D2. The remaining duty cycle D3=1−D1−D2 is the duty cycle of zerostate S3, and should be split into two parts to be inserted between D1and D2. In steady state, the ideal output voltages are:

${{V\; 1} = {\frac{Vin}{2D\; 1}\frac{n\; 2}{n\; 1}}},{{V\; 2} = {2V\; 1}},{{V\; 3} = {4V\; 1}},{{V\; 4} = {8V\; 1}}$

Therefore, by controlling D1, the output voltages can be adjusted to thedesired value. This control can be implemented as a current mode orvoltage mode closed-loop control if precise regulation is required.

To maintain the doubling relationship between V2 and V1, anotherclosed-loop control may be used to adjust the relative value of V2. IfV2 is higher than the ideal value (2*V1), D2 should be increased. If V2is lower than the ideal value, D2 should be reduced.

If the duty cycle of State 3 (zero state) is intentionally kept close tozero, then this converter becomes a multi-output bus converter, whichmay have high efficiency.

Because the primary side switches are not clamped, there may be highvoltage stress during a switch's turn-off. Proper measures, such as asnubber or an active clamp circuit can be used to reduce the voltagestress. In FIG. 19, a snubber/clamp circuit 194 is shown. In a preferredembodiment, the circuit 194 may be a passive snubber consisting of acapacitor and a resistor. In another embodiment, the circuit 194 mayhave one or more active clamps.

FIG. 20 shows the simulated key waveforms of the converter shown in FIG.19 in an example implementation, and verifies the operation principlediscussed above. Although doubling relationship is shown in thisembodiment for the voltages in a pair, other relationships are possibleby changing the relationship between D1 and D2. If D1=D2, then the twovoltages in pair 191 or 192 will be equal.

In some power applications such as smart phones and other mobiledevices, there is no isolation needed between the input and the outputsof the power supply. Then a buck converter based multi-outputconfiguration, with similar principles as discussed before, can be usedto generate the input voltages for the power synthesizer. FIG. 21 showstwo more examples. In FIG. 21(a), flyback windings n2, n3 and n4 areadded to the magnetic component (inductor T1), and S2 through S4 work assynchronous rectifiers (in rectifier networks 181 and 182). When S1conducts (S1 d is gated off) in the buck switch network 180, switches inthe rectifier networks 181 and 182 (S2 through S4) are gated off. WhenSd1 conducts (S1 is turned off) in the freewheeling mode of the buckconverter, S2 through S4 are gated on so power is transferred from V1 toV2 through V4, and the relationship between the output voltages isdetermined by the turns-ratios of the windings. The circuit in FIG.21(b) works similarly to the circuit in FIG. 21(a), but the rectifiernetworks 182 and 181 are arranged slightly differently.

The power synthesizer described above has a minimum voltage of zero. Ifthe power amplifier needs a higher minimum voltage, another voltagesource can be added to the power synthesizer. FIG. 22 shows an exampleof a fast dynamic power supply based on a buck-based topology. V1 is theminimum voltage applied to the PA and is generated by the buck switchcell 180 directly. Through using windings n2, n3 and n4 with propernumber of turns, V2, V3 and V4 can have the correct voltages, and arefed into an eight-level power synthesizer consisting of a two-level cell140 and 1 four-level cell 150. With this arrangement, eight differentvoltage levels can be achieved.

Please note that in all the above descriptions, the input power sourceVin can be a primary power supply such as a battery pack, a batteryplant, a photovoltaic cell, a fuel cell, or the output of a powerconverter such as a dc-dc converter, or an ac-dc converter. When a dc-dcconverter or an ac-dc converter is used, the input voltage Vin can bedynamically adjusted. The input voltages to the power synthesizer (V1,V2, and so on) can be adjusted dynamically through changing Vin orcontrolling the switches in the multi-output power supply to optimizethe performance (such as the efficiency) of the whole system. Forexample, when the system transfer power is increased due to heavierusage of users in a base station, the voltages can be increased. Whenthe needed transfer power is reduced, the voltages can be decreasedaccordingly. For ET systems, the output voltages of the power supply canbe controlled with a relatively slow voltage control loop (compared tothe frequency of power synthesizer, or the envelope spectrum of thesignal processed by the power amplifier) and follow the envelope peakswhich are related to the peak or average power of the power amplifier'soutput over a duration of time, and change relatively slowly. Such asystem is sometimes called average mode ET control.

Please note also that although in this disclosure multi outputs are usedto describe the power supply topologies and control schemes, theconcepts and principles can also be used for single output converterswith straightforward modification (such as remove unused secondarywindings and their associated circuitry). Of course, single-output powersupply may not need the double-side resonance technique if the outputvoltage is sensed directly at the output. However, sensing the outputvoltage through a feedback winding at the primary side can still workwell with double-side resonance technique even for the single-outputapplications, and using the feedback control to automatically operate aresonant converter at or near its resonant frequency is still a veryeffective technique for single-output converters.

One aspect to improve the performance of a power converter is to limitthe range of the switching frequency. This is very important especiallyin resonant power converters and resonant power supplies. FIG. 23 showsa technique to accomplish this purpose. In an embodiment of a resonantpower converter, an input voltage circuit 155 generates an input voltageVin. The circuit 155 may be an acdc power converter connected to an acpower source, or a power terminal or power bus to bring in an externaldc source. Vin may vary in a wide range due to various reasons, such asthe voltage ripple in the output voltage of an acdc converter or dioderectifier. A switch network 180 converts Vin into an ac or pulsewaveform, which is fed into a primary resonant tank 183. Optionaltransformer and secondary resonant tank 185 are coupled to the primaryresonant tank 183 and output circuit 182, which generates one or moreoutput voltages or currents. A frequency control block 161 controls thefrequency of the switch network 180, and thus controls the operation ofthe resonant power converter 2300 and may regulate its output voltage,current or power. Resonant tanks 183 and/or 185 may have more than oneresonant modes. For example, one of a resonant capacitor or inductor ina resonant tank may be variable and take different values, so that theresonant frequency and/or characteristic impedance of the resonant tankcan change in a certain range, creating different resonant modes withdifferent operating characteristics in the operation of the resonantpower converter. A resonant mode selection circuit 162 is used to selecta resonant mode according to the operating conditions and requirements.For example, one objective of the resonant mode selection block 162 maybe to limit the switching frequency range of the resonant powerconverter so its design can be optimized. To do this, block 162 may usethe input voltage as an input to choose a suitable resonant mode for agiven range of input voltage. If the input voltage moves out of thisrange then a different resonant mode is chosen which is more suitablefor the changed input voltage. Alternatively, block 162 may useinformation from the frequency control block 161 as an input. When thefrequency control requires the switching frequency of the resonant powerconverter to enter a undesired range (for example, to go higher than aupper limit or lower than a lower limit), the resonant mode selectionblock 162 changes the resonant mode so the switching frequency willremain in the desired range. In this way, even if Vin may change in awide range due to a line-frequency ripple or other reasons, or anoperating characteristics may change in a wide range which may cause theswitching frequency to change in a wide range, by switching amongdifferent resonant modes properly the resonant mode selection block 162can maintain the operating of the resonant converter within a narrowerswitching frequency range, which may result in lower cost design and/orbetter performance of the resonant converter. Of course, the resonantmode selection block 162 may have other objectives, such as to increasethe efficiency of the resonant power converter. This switched resonantmode operation can be implemented in primary resonant tank 183,secondary resonant tank 185, or both.

FIG. 24 shows an example to implement the technique of switched resonantmode described above. Sp1 and sp2 form a half-bridge switch network 180.Secondary circuit 186 is not shown in detail for the sake of brevity,and may consist of any suitable circuit to produce one or more suitableoutputs. Lrp, which may include the leakage inductance of transformer T,is a resonant inductor. Cx0, Cp1 and Cp2, as well as the switchablecapacitor network 163 consisting of Cx1, Cx2, Sx1 and Sx2, formequivalently a resonant capacitor. By controlling the ON/OFF states ofSx1 and Sx2, four different capacitances, including 0, Cx1, Cx2, andCx1+Cx2, can be put into parallel with Cx0, resulting in 4 differentresonant capacitances for the resonant tank 183. This is equivalent tochange the capacitance of the equivalent resonant capacitor. In thisway, 4 resonant modes are generated, and can be switched by resonantmode selection block 162.

In low power application, a single-end topology such as a flyback,forward, or class-E resonant converter may be more desired due to lowerimplementation cost. FIG. 25 shows a single-end resonant topology whichis a hybrid of a flyback converter and a class-E converter. Sp1 forms aprimary switch network 180, Cp1 and the primary winding of transformer T(184) form a primary switch network 183, Cs1 and a secondary winding oftransformer T form a secondary resonant tank 185. Cp1 may include theoutput capacitance of power switch Sp1. The transformer T may beimplemented on a PCB (printed circuit board) or multiple PCBs to haverelatively consistent magnetizing inductance and leakage inductance.Multiple PCBs may be separated to increase safety distance or for othersystem consideration. Of course, separate inductors may be added to theprimary winding or secondary winding of the transformer, and differentresonant tank configurations, such as a parallel resonant orparallel-series resonant tank, may be used for the primary or secondaryresonant tank. A half-bridge switch network of S11 and S12 and an outputcapacitor Co form an output circuit 182 to produce an output Vo. Ifdesired, a full-bridge switch network may be used in the output circuit.Furthermore, multiple secondary resonant circuits and output circuitsmay be used to produce multiple outputs if needed. Through proper dutycycle and/or switching frequency control, Sp1 can work in zero-voltageswitching, and the converter can output an output voltage in a suitablerange.

The switched resonant mode operation can be applied to this resonantconverter by making Cp1, Cs1, or both changeable. FIG. 26 shows animplementation example to change the equivalent capacitance of Cp1. Inthe embodiment of FIG. 26, a switchable capacitor network 163 consistingof Cx1, Cx2, Sx1 and Sx2 is put in parallel with Cp1. A resonant modeselection block 162 can use Vin or frequency information in frequencycontrol block 161 to decide the status of Sx1 and Sx2, and thus choosingthe right resonant mode for the primary resonant tank, so a goodperformance can be obtained with a low implementation cost. Please notethat the capacitance of Cs1 in the secondary resonant tank 185 may alsobe changed dynamically. To reduce cost, it's better to adjust Cs1 justaccording to a variable in the secondary circuit, such as an outputvoltage change or load change.

With the switching frequency range of a resonant converter reduced bythe switched resonant mode technique, resonant gate drivers such as theones described in FIGS. 15(a), 15(b), 15(c) and 15(d) can be more easilyused. For a single-end topology such as the one shown in FIG. 26, asimpler resonant gate drive topology shown in FIG. 27 may be used. Inthis embodiment, switch network 180 consisting of switches S1 and S2converts a dc drive voltage Vcc into a pulse waveform, and couples thewaveforms to a resonant tank 183, which consists of a resonant inductorLg, and a resonant capacitor Cgs. Lg may consist of a parasiticinductance in the gate path, such as PCB trace inductance and/or bondwire inductance coupled between the switch network 180 and the gate andsource of the main power switch, as well as any added discrete inductor.Cgs may include parasitic capacitance such as any PCB trace or Padcapacitance and the input capacitance of the main power switch (notshown in the figure) which is driven by this gate drive circuit, and mayinclude any added discrete capacitor. Switch S3 is optional and mayclamp the gate-source voltage and ensure the turn-off of the main powerswitch, as will be discussed later. FIG. 28 shows a phase diagram of anintended operation of the gate drive circuit in FIG. 27. Similar to thedrive circuit shown in FIG. 15(a) and the phase diagram in FIG. 15(b),the resonant gate drive circuit shown in FIG. 27 may utilize the conceptof composite driving technique where the resonant tank in the gate drivecircuit may operate in more than one states in the charging phase anddischarging phase of a switching cycle.

The gate drive circuit's operation can be explained more clearly withthe help of the phase diagram shown in FIG. 28. We can investigate firstthe operation of the circuit without S3. Assuming initially the circuitis at the following condition:

Vgs=0,Ig≈Vcc/Zo, where Zo is the characteristic impedance of theresonant tank, i.e. the square root of (Lg/Cgs).

This corresponds to point A in FIG. 28. To initiate a turn-on processfor the main switch, S2 should be turned off, and S1 should be turnedon. Then the state variable (Vgs, Ig) of the resonant tank 183 will moveclockwise along the arc GAB in the phase diagram. The arc GAB is part ofa circuit (the resonant circle) with its center at (0, 0) and a radiusof about Vcc. This is a resonant state, and during this state, Vgsgradually increases, while Ig gradually decreases, both in a sinusoidalfashion. When Vgs exceeds the gate threshold voltage of the main powerswitch, the main power switch will turn on. When Vgs approaches Vcc, forexample Vgs reaches a level of 90% of Vcc, S1 is turned off and S2 isturned on. This corresponds to point B in FIG. 28. The duration of theresonant state is a little lower than ¼ of the resonant perioddetermined by the values of Lg and Cgs. After this, the state variable(Vgs, Ig) of resonant tank 183 will enter a small arc BCD or a smallcircle BCDEB in the phase diagram, with the center of the both at (+Vcc,0). Because the radius of the small circle or small arc is much smallerthan Vcc, during this state Vgs fluctuates around Vcc slightly duringthis mode (for example staying within +/−30% of the average value duringthis mode), and Ig fluctuates around zero slightly, so roughly the gatevoltage of the main power switch can be considered to be pseudo“clamped” around Vcc with a small ripple, which can be designed to beless than a limit suitable for driving the power switch, say within 30%of the average voltage during this state. This state is a pseudo clampedstate. Each circle corresponds to a resonant period in time. Because inthis state the state variable can circle the center (Vcc, 0) as manytimes as required, the gate drive voltage can stay at a level suitablefor driving a switch in the pseudo “clamped” mode for a relatively longand controllable time, which is terminated by the start of the turn-offphase at point D. Therefore, by adjusting the duration of the pseudoclamped mode, the on time of the main switch can be controlled, and thusthe duty cycle and/or the switching frequency of the power converter canbe controlled. Point D is a mirror to Point B along Vgs axis, at whichVgs is the same as at Point B, but Ig changes direction compared to atpoint B. At this point S2 is turned off and S1 is turned on, so thestate variable of the resonant tank 183 will moves clockwise along thearc DFGA in the phase diagram shown in FIG. 28. Ideally, the arc DFGA ispart of the resonant circle with its center at (0, 0) and a radius ofabout Vcc. During the first part of this resonant state, Vgs graduallydecreases, while Ig gradually increases, both in a sinusoidal fashion.When Vgs moves below the gate threshold voltage of the main powerswitch, the main power switch will turn off After this, the statevariable of the resonant tank 183 continues to move along the arc DFGA,with Vgs and Ig both changes polarity along the course, and reachespoint A at the end. The time duration of the resonant state from point Dto point A is about ¾ of a resonant period. Then next switching cyclebegins repeating the above process.

With the circuit in FIG. 27 and the phase diagram in FIG. 28, theduration of main power switch's OFF state is relatively fixed. Byadjusting the relative duration of the pseudo clamp mode, the drivecircuit's operation is more independent of the resonant tank parameters,and the transition time between a switch's states can be faster than intraditional control. Moreover, the duty cycle of the main switch caneven be adjusted in certain degree. A potential shortcoming is that themain switch's turn-off time and duration may not be fully controlled. Toimprove this, S3 can be add as shown in the dotted line in FIG. 27. Theoperation of the circuit with S3 connected is basically the same duringthe turn-on process and the first part of the turn-off transition of themain power switch, during which S3 is turned off. The difference startswhen the state variable of the resonant tank 183 reaches point F. Atpoint F, due to conduction of the body diode or anti-parallel diode ofS3, the state trajectory will stop movement, so Vgs is clamped at zero,with Ig equal to −Vcc/Zo. This is a first gate freewheeling state,during which S3 may be gated on to reduce the power loss. Ignoring theconduction loss of the drive current path, this state can be as long asforever. In a practical operation, even with the conduction lossconsidered this state can last a long enough time to allow a good dutycycle control of most power converters. When it is desired to start aturn-on process for the main power switch, S1 can be used off and S2 canbe turned on possible. Please note that S2 may be turned on withzero-voltage as the negative Ig can force the body diode of S2 toconduct before S2 is gated on. During this state of the operation, thestate variable of the resonant tank 183 moves along the dotted straightline FOA in the phase diagram. During this state, Vgs is clamed to zeroby S3, but Ig is charged linearly by Vcc, so this stage can be calledinductor charging stage. If S3 was not gated on previously, it can begated on before Ig reaches zero. Also, during this stage if S2 is turnedoff and S1 is turned on, the gate drive circuit can enter into secondgate freewheeling stage in which Vgs is clamped zero, and Ig doesn'tchange. If Ig is close to zero, then the power loss in the gate drivecircuit is very small in this state. Therefore, it may be desirable touse this second gate freewheeling state alone or in combination with thefirst gate freewheeling state to control the turn-off time of the mainswitch. When the state variable reaches point A, S2 and S3 can be turnedoff, and S1 can be turned on. The positive Ig makes it possible to turnon S1 with zero-voltage. After this, the resonant gate drive circuitenters the resonant stage in the turn-on process as discussed above.

At any time the main switch needs to be turned off such as in a faultycondition or shut-down operation, S1 and S2 should be turned off and S3can be turned on. In this way, the turn-on of S3 can ensure the turn-offof the main power switch. Similar turn-off mechanism can be applied tothe gate drive circuit shown in FIG. 15(c), where with proper circuitparameters, Sg3 in secondary circuit block 186 can reduce the outputgate voltage of Drive 3 to close to zero when the switch network 180produces zero voltage, i.e. when both S1 and S4 (or alternatively bothS2 and S3) are turned on. If similar circuit is used for other drives,all main power switches may have a long turn-off state, which allow apower converter to be in a shut-down or fault-protection mode.

Another consideration is the start-up of a resonant gate drive circuit.It is interesting to know that the state trajectory during the resonantstates in both the phase diagrams shown in FIG. 15(b) and FIG. 28 ispart of a resonant circle with its center at (0, 0) and a radius ofabout Vcc. Since the initial condition of the state variables of theresonant gate drive circuit is (0, 0), the gate-drive start-up can startwith Vcc being applied to the resonant tank in the circuit. In thecircuits shown in FIGS. 15(a) and 15(c), it means turn-on of S2 and S4while turn-off of S1 and S3. In the circuit shown in FIG. 27, it meansturn-on of S2 and turn-off of S1 and S3 (if it is connected). This willforce the state variables of the gate drive resonant tank to move a longa start-up circle with its center at (Vcc, 0) and a radius of Vcc in thephase diagram. This start-up circle will cross the resonant circle afterabout ⅛ resonant period. At the cross point, Vgs is about 70% of Vcc.Therefore, the cross point can be determined by timing the duration ofstart-up state, or by comparing Vgs to a value close to 70% of Vcc. Atthe cross over point, the gate drive start-up process is over, and thestatus of switches in a gate drive circuit can be changed to thecorresponding switch status in the corresponding resonant state asdiscussed above. According to this gate-drive start-up scheme, it onlyneeds about ⅛ resonant period of the drive resonant tank to bring thegate drive circuits into normal operation. Once the gate drive circuitcan start normal operation, the switch frequency and/or duty cycle ofmain power switches should be changed to reduce voltage and currentstresses in a soft-start process, as discussed previously.

The key to the above drive schemes is to accomplish correct controltiming for the drive switches S1 through S4. These switches' ON and OFFinstants can be timed according to the duration of each state asdiscussed previously. However, when the resonant inductor and resonantcapacitor, and thus the resonant period, have relatively big tolerance,it is difficult to do such fixed timing without careful measurement orcalibration of the resonant elements or resonant period. Alternatively,the correct timing may be determined by comparing the resonant capacitorvoltage to given reference points (such as a value close to +Vcc or−Vcc), and/or comparing the resonant inductor current to given referencepoints, such as zero or Vcc/Zo, so this is basically a drive scheme withfeedback. The measurement of capacitor voltage is relativelystraight-forward. The measurement of resonant inductor current (such asIg) can be done on a resistor in the gate drive path, or through currentmirrors of the gate drive switches. With modern analog and digitaltechniques, the signal sensing and processing according to the aboveschemes are all possible, and the sensitive part of a gate drive circuitmay be implemented on an integrated circuit.

This resonant gate drive circuit is actually a resonant power converter.The circuit and technique discussed here can also be used in resonantpower converters with straightforward modifications to suit the outputrequirements of the converter.

Because the dynamic power supply has high bandwidth requirements, thepower synthesizer needs to work at very high frequency. Powerintegration techniques can be used to improve efficiency and reducenoise whenever possible. The switches are arranged in totem poleconfigurations in the synthesizer. In one embodiment, the two switchesand their drive circuit in a totem pole are integrated together in asingle die or a single package. In another embodiment, the four switchesin a basic four-level cell and their drive circuits are integrated in asingle die or package. In other embodiments, all switches of a powersynthesizer and their gate drive circuit are better to be integratedinto a single die, or into a single package. The control circuit of thesynthesizer can be integrated too. It's preferable to include somebypass capacitors, such as C2, C3 and C4 in FIG. 21 in the same packageor the same die as the switches. If possible, the power supply switches(for example S1 through S4 and S1 d in FIG. 21) can also be integratedinto the same die or the same package. The drive circuit and controlcircuit of the multi-output power supply can be also integrated into thedie or the package. In an embodiment, the active components and keypassive components of the whole dynamic power supply can be integratedinto a package, with one or more semiconductor dies and optionaldiscrete passive components. Finally, the power amplifier itself can beintegrated with the power solution in a single die, or in a multi-chipmodule package. In this way, the effect of package parasitics can beminimized, and optimum performance can be achieved.

The above discussion of the advanced power solutions is made mainly inthe context of power amplifier applications, since power amplifiers area very suitable target for high performance power technologies. However,the applications of the power technologies discussed are not limited topower amplifiers, and can be used in any other applications, devices andequipment which require high efficiency power conversion and powercontrol. The power technologies can be used in stand-alone powerdevices, power chips, power modules and power supplies systems, or beused in devices or systems with power solutions as a built-in function.

Although embodiments of the present invention and its advantages havebeen described in detail, it should be understood that various changes,substitutions and alterations can be made herein without departing fromthe spirit and scope of the invention as defined by the appended claims.

Moreover, the scope of the present application is not intended to belimited to the particular embodiments of the process, machine,manufacture, composition of matter, means, methods and steps describedin the specification. As one of ordinary skill in the art will readilyappreciate from the disclosure of the present invention, processes,machines, manufacture, compositions of matter, means, methods, or steps,presently existing or later to be developed, that perform substantiallythe same function or achieve substantially the same result as thecorresponding embodiments described herein may be utilized according tothe present invention. Accordingly, the appended claims are intended toinclude within their scope such processes, machines, manufacture,compositions of matter, means, methods, or steps.

What is claimed is:
 1. A system comprising: an input port having aninput voltage; an output port having an output voltage; a powerconverter comprising a switch network having a plurality of powerswitches and a first resonant tank comprising a first resonant capacitorand a first resonant inductor, wherein at least one resonant componentwithin the first resonant capacitor and the first resonant inductor is aswitchable component configured to switch between different values; aresonant mode selection block configured to adjust a value of theswitchable component to maintain a performance of the system; and acontroller configured to adjust a switching frequency or a duty cycle ofthe power converter.
 2. The system of claim 1, wherein the resonant modeselection block adjusts a value of the switchable component in responseto a change of the input voltage.
 3. The system of claim 1, wherein theresonant mode selection block adjusts a value of the resonant componentin response to a change of the output voltage.
 4. The system of claim 1,wherein the controller adjusts a value of the resonant component inresponse to a change of the switching frequency.
 5. The system of claim4, wherein the switching frequency is limited within a preset range. 6.The system of claim 1, wherein a primary winding of a power transformeris connected between the input port and a power switch, and wherein thefirst resonant tank comprises an inductance of the power transformer anda parasitic capacitance of a power switch in the switch network.
 7. Thesystem of claim 6, wherein the first resonant tank comprises a capacitorin parallel with the switch, and wherein the capacitor is the switchablecomponent.
 8. The system of claim 6, further comprising a secondresonant tank coupled between the output port and a secondary winding ofthe power transformer, wherein the second resonant tank comprises aninductance of the power transformer.
 9. The system of claim 1, whereinthe resonant mode selection block is configured to adjust a value of theswitchable component in response to a fault of the system.
 10. Thesystem of claim 1, wherein the switchable component comprises aplurality of switch-capacitor branches, and wherein a switch-capacitorbranch comprises a capacitor and a switch.
 11. An apparatus comprising:an input port having an input voltage; an output port having an outputvoltage; a transformer having a primary winding and a secondarywingding, wherein the primary wingding is coupled to the input port; apower switch coupled between the primary winding and the input port; arectifier circuit coupled between the secondary winding and the outputport; a primary resonant tank comprising: a primary resonant capacitorin parallel with the power switch; and a primary resonant inductorcomprising the primary winding; a secondary resonant tank comprising: asecondary resonant capacitor coupled between the secondary winding andthe rectifier circuit; and a secondary resonant inductor comprising thesecondary winding; and a control block configured to control at leastone of a duty cycle and a switching frequency of the power switch. 12.The apparatus of claim 11, wherein one of the first resonant capacitorand the second resonant capacitor is a switchable device configured tohave different capacitance values in response to a change in the inputvoltage or in the output voltage.
 13. The apparatus of claim 12, whereinthe switchable device comprises a plurality of switch-capacitor branchesconfigured to control an operational parameter of the apparatus, andwherein a switch-capacitor branch comprises a capacitor and a switch.14. The apparatus of claim 13, wherein the switchable devices isconfigured to limit a switching frequency of the power switch within apreset range.
 15. The apparatus of claim 13, wherein the switchabledevices is configured to protect the apparatus against a fault.
 16. Theapparatus of claim 11, wherein the duty cycle of the switch iscontrolled to regulate the output voltage.
 17. A method comprising:configuring a power converter comprising an input port having an inputvoltage, an output port having an output voltage, a switch networkhaving a plurality of power switches, and a resonant tank comprising aresonant capacitor and a resonant inductor, wherein at least oneresonant component within the resonant capacitor and the resonantinductor is a switchable component configured to switch betweendifferent values; configuring a resonant mode selection block to adjusta value of the switchable component to maintain a performance of thesystem; and configuring a controller to adjust a switching frequency ora duty cycle of the power converter.
 18. The method of claim 17, furthercomprising adjusting a value of the switchable component in response toa change of the input voltage.
 19. The system of claim 17, furthercomprising adjusting a value of the resonant component in response to achange of the output voltage.
 20. The system of claim 17, furthercomprising adjusting a value of the resonant component in response to achange of the switching frequency.